FIG. 1 is a high level block diagram of a switching resonant voltage converter. Among resonant voltage converters having the basic architecture of FIG. 1, and that are classified based upon the configuration of the resonant circuit that is used, there is the LLC resonant voltage converter. A half-bridge driven architecture of such a converter is illustrated in FIG. 2.
For simplicity, reference will be made to half-bridge driven voltage converters, though the addressed technical problems also affect full-bridge driven voltage converters. One of the weak points of this architecture, especially when functioning at high power levels (>1 kW), is tied to the AC current that flows through the output capacitor COUT. This AC current has large peak and rms values that may require the use of a larger, and thus more encumbering bank of capacitors for the capacitance COUT than for a forward voltage converter of the same output voltage and power. This considerably burdens the LLC resonant converter, especially in power applications of relatively large power density such as, for example, power supply systems for servers or for telecommunication systems, in which its high efficiency characteristic is particularly advantageous.
The multi-phase or “interleaving” techniques may prevent this drawback. A multi-phase voltage converter may be obtained by connecting in parallel two or more switching converters of a same architecture to make them share the same input voltage generator, and supply the same output load. Moreover, with an appropriate phase control of the driving signals of the power switches, it may be possible to minimize or, in certain cases, even to practically nullify the ripple on the output current (sometimes even on the input current) of the converters.
Other advantages of the multi-phase approach are the possibility of subdividing the overall power requirement among a number of smaller converters thus making a larger power density possible and optimizing efficiency over a larger interval of load currents using the “phase shedding” technique. That is, turning off one or more phases when the load decreases, and managing the reduced requirement with a reduced number of converters, thus reducing losses due to parasitic components of the power circuits that may become dominant with low power conditions.
The interleaving technique achieves:
1) reduction of the ripple of the output and input currents of the converter;
2) reduction of the power managed by each converter with a consequent optimization of their dimensioning;
3) increased efficiency over a wide range of output load because of the turning off one or more phase circuits when functioning at low power and reduction of losses due to parasitic components; and
4) greater power density and smaller form factor. To achieve the above beneficial effects, it may be essential to ensure that the load of the converter be subdivided as equally as possible among the phase circuits. This is a serious obstacle to implementation of “interleaving” techniques in multi-phase resonant voltage converters.
To better illustrate the problem, reference is made to the three-phase LLC resonant voltage converter of FIG. 4, though the same considerations hold for resonant converters of a different type and with any number of phases. The distinct phase circuits are driven at the same frequency, and the driving signals of the power switches are mutually phased apart by 120° making the currents of the output diodes superpose with continuity. This functioning condition is illustrated in the time graphs of FIG. 5.
In a first harmonic approximation, the functioning of a single LLC resonant phase is quantitatively described by means of characteristic curves, as the ones depicted in FIG. 6. In these figures the abscissa is the operating frequency x, normalized to the series resonant frequency associated with the elements Cr and Ls of the resonant circuit of FIG. 2, and the ordinate is the ratio M between the voltage on the nodes of the secondary winding, which is equal to the sum of the output voltage and the voltage drop on the secondary rectifiers translated to the primary circuit, and the input voltage. Each characteristic curve is associated to the quality factor Q of the resonant circuit that is inversely proportional to the output resistance ROUT. As a consequence, Q is substantially proportional to the output current IOUT, and each curve is substantially associated with a value of the load current.
The three phase circuits are powered with the same input voltage, they “see” the same output voltage, and work at the same frequency. If the three phase circuits are exactly identical among them, they will work with the same current amplitude, as shown in FIG. 5.
Nevertheless, in a real world implementation, the inevitable tolerances of the components must be taken into consideration. Thus, the three phase circuits may have different values of the ratio M because of the effect of different voltage drops on the respective secondary rectifiers and of different values of x, and/or of the proportionality constant between Q and IOUT because of differences among the values of Ls, Cr and Lp of the three resonant circuits. As a consequence, the currents in the various phase circuits will differ, and one of them may even provide the whole power for the load, while the other phases may be inactive.
These theoretical predictions are confirmed by simulation. In the diagrams of FIG. 7 the same signals of FIG. 5 relating to the converter of FIG. 4 are shown, but the value of the capacitor Cr of the phase circuit 2 is reduced by 10% and that of phase circuit 3 is increased by 10%. The currents through phase circuits 1 and 3 are close to zero, and the current of phase circuit 2 is almost equal to the output current. Solely phase circuit 2 is effectively working, and there is no interleaving among the phase circuits. More precisely, compared with the ideal case of FIG. 5, the average output current of phase circuit 1 is reduced by 97.4%, that of phase circuit 2 is increased by 297%, and that of phase circuit 3 is zero; the peak-to-peak amplitude of the ripple of the output current, divided by its mean value, has changed from 17.8% to 165%. The rms value of the output current divided by the mean value is 114%. The rms value of the AC component is 55% of the mean value. As could have been expected, these values resemble those of a single phase LLC resonant voltage converter. This situation, verified in an exemplary test case, is unacceptable because it would force to size each phase converter for delivering the whole output power, without any reduction of the ripple of the output current.
Published U.S. patent application No. 2008/0298093 A1 “Multi-phase resonant converters for DC-DC application,” discloses a three-phase LLC resonant voltage converter including three half-bridges connected to the same input bus (re.: the architecture of FIG. 4, in which a further phase circuit in parallel to the two depicted phase circuits has been added), and shows that it is possible to balance the phase currents. Indeed, only the ideal case of exactly identical converters is considered, neglecting spreads among the components.
In U.S. Pat. No. 6,970,366, entitled “Phase-shifted resonant converter having reduced output ripple”, a system of two LLC resonant converters, synchronized and mutually phased apart by 90° to minimize the overall ripple is disclosed. The document is silent about balancing the two phases.
In the article by H. Figge et al., entitled “Paralleling of LLC resonant converter using frequency controlled current balancing”, IEEE PESC 2008, June 2008, pp. 1080-1085, a system is disclosed in which a DC-DC buck conversion stage is installed upstream of a two-phases LLC resonant converter. The regulation loop of the output voltage modulates the voltage generated by the buck (and, thus, the input voltage of the two half-bridges). A regulation loop of the balancing of the currents through the two phases determines the switching frequency of the half-bridges that are relatively phased apart by 90°. This architecture addresses the problem of balancing the currents at the cost of employing an additional conversion stage that reduces overall efficiency and increases the overall complexity of the converter circuit.
The degree of freedom to balance the currents could be provided by duty-cycle adjustment. In this way, the mean value of the voltage applied to each phase would be adjustable. Nevertheless, as shown in the simulations of FIG. 8, this approach may be followed only if small adjustments are sufficient for obtaining a satisfactory balancing. Indeed, a duty-cycle significantly different from 50% would generate strongly asymmetrical currents in the secondary windings of the transformer and in the output diodes, thus the balancing problem would be merely shifted elsewhere. For implementing this method, the reactive components of the resonating circuits would have to be selected accurately, which is costly.
The recognized problems of known interleaved resonant architectures could be resumed, because of their marked sensitivity to differences among the power circuits and difficulty of finding a control variable that would be conveniently used for compensating the consequent unbalancing of the currents among the single phase circuits. This is an indispensable condition for reducing the ripple of the output current, the main reason for implementing the interleaving.